Clock and data recovery devices with fractional-N PLL

ABSTRACT

The present invention relates to data communication and electrical circuits. More specifically, embodiments of the present invention provide a clock and data recovery (CDR) architecture implementation for high data rate wireline communication links. In an embodiment, a CDR device includes a phase detector, a loop filter, and a fractional-N PLL. The fractional-N PLL generates output clock signal based on output of the loop filter. There are other embodiments as well.

CROSS-REFERENCES TO RELATED APPLICATIONS

The present invention is a continuation application to U.S. patent application Ser. No. 16/127,103, filed on 10 Sep. 2018, entitled “CLOCK AND DATA RECOVERY DEVICES WITH FRACTIONAL-N PLL”, which is incorporated by reference herein for all purposes.

STATEMENT AS TO RIGHTS TO INVENTIONS MADE UNDER FEDERALLY SPONSORED RESEARCH AND DEVELOPMENT

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BACKGROUND OF THE INVENTION

The present invention relates to data communication and electrical circuits.

Over the last few decades, the use of communication networks has exploded. In the early days of the Internet, popular applications were limited to emails, bulletin board, and mostly informational and text-based web page surfing, and the amount of data transferred was usually relatively small. Today, Internet and mobile applications demand a huge amount of bandwidth for transferring photo, video, music, and other multimedia files. For example, a social network like Facebook processes more than 500 TB of data daily.

Clock and data recovery (CDR) devices are used in a wide range of applications. For example, to process data received over a communication network, a receiver relies on its CDR to generate a clock signal based on the received data. The performance of the receiver relies on the performance of its CDR and other components. Over the time, there have been many different types of CDR designs and implementations, however, they have been inadequate for the reasons explained below. Therefore, new and improved CDR devices are desired.

BRIEF SUMMARY OF THE INVENTION

The present invention relates to data communication and electrical circuits. More specifically, embodiments of the present invention provide a clock and data recovery (CDR) architecture implementation for high data rate wireline communication links. In an embodiment, a CDR device includes a phase detector, a loop filter, and a fractional-N PLL. The fractional-N PLL generates output clock signal based on output of the loop filter. There are other embodiments as well.

According to an embodiment, the present invention provides A clock data recovery (CDR) device, which includes a first phase detection module configured to receive digital input signal and determine a difference between phases of the digital input signal and an output clock signal. The device additionally includes a loop filter module coupled to the phase detector module and configured to generate a frequency control word. The device also includes a fractional-N phase lock loop (Frac-N PLL) configured to generate the output clock signal based on the frequency control word. The Frac-N PLL includes a second phase detection module coupled to a reference clock signal. The Frac-N PLL also includes a charge pump coupled to the second phase detection module. The Frac-N PLL additionally includes an analog loop filter coupled to the charge pump. The Frac-N PLL also includes a sigma delta modulator configured to process the frequency control word.

According to another embodiment, the present invention provides a clock data recovery (CDR) device that includes a first phase detection module configured to receive digital input signal and determine a difference between phases of the digital input signal and an output clock signal. The device also includes a loop filter module coupled to the phase detector module and configured to generate a frequency control word. The device additionally includes a fractional-N phase lock loop (Frac-N PLL) configured to generate the output clock signal based on the frequency control word, the Frac-N PLL comprising a high-pass filter for processing the frequency control word. The high-pass filter has an inverse filter and an anti-aliasing filter.

According to yet another embodiment, the present invention provides a clock data recovery (CDR) device, which has a first phase detection module configured to receive digital input signal and determine a difference between the digital input signal and an output clock signal. The device also includes a loop filter module coupled to the phase detector module and configured to generate a frequency control word. The device additionally includes a fractional-N phase lock loop (Frac-N PLL) configured to generate the output clock signal based on the frequency control word. The Frac-N PLL has a voltage controlled oscillator configured on a feedforward signal path and a current mode digital-to-analog converter.

It is to be appreciated that embodiments of the present invention provide many advantages over conventional techniques. Among other things, CDR devices implemented according to embodiments of the present invention are smaller and more efficient compared to existing devices (e.g., phase-interpolator or digitally controlled oscillator implementations). For example, compared to CDRs with digitally controlled oscillators, CDRs with fractional-N PLL have better immunity to various types of coupling noise. Additionally, a fractional-N PLL based CDR can provide independent gain suitable for the CDR device.

Embodiments of the present invention can be implemented in conjunction with existing systems and processes. For example, CDR devices implemented with Frac-N PLLs according to the present invention can be used for a wide range of applications and are compatible with existing systems and architectures. Additionally, CDR devices according to the present invention can be manufactured using existing manufacturing processes and equipment. There are other benefits as well.

The present invention achieves these benefits and others in the context of known technology. However, a further understanding of the nature and advantages of the present invention may be realized by reference to the latter portions of the specification and attached drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The following diagrams are merely examples, which should not unduly limit the scope of the claims herein. One of ordinary skill in the art would recognize many other variations, modifications, and alternatives. It is also understood that the examples and embodiments described herein are for illustrative purposes only and that various modifications or changes in light thereof will be suggested to persons skilled in the art and are to be included within the spirit and purview of this process and scope of the appended claims.

FIG. 1 is a simplified diagram illustrating a fractional-N PLL according to an embodiment of the present invention.

FIG. 2 provides a small signal model of a multi-modulus divider according to embodiments of the present invention.

FIG. 3 is a simplified diagram illustrating a digitally controlled oscillator (DCO) based CDR loop.

FIG. 4 is a simplified diagram illustrating a Frac-N PLL incorporated inside a CDR loop according to embodiments of the present invention.

FIG. 5 is a simplified diagram illustrating small signal phase domain model of a CDR loop with a DCO.

FIG. 6 is a simplified diagram illustrating a small signal model of Frac-N PLL functioning as a DCO according to embodiments of the present invention.

FIG. 7 is a plot illustrating CDR error transfer functions comparing ideal DCO and actual DCO implementation (e.g., with Frac-N PLL) with a pole that contributes to undesired error transfer function peaking.

FIG. 8 provides plots illustrating CDR error transfer function peaking (top) and Frac-N PLL jitter (bottom) as function of Frac-N PLL bandwidth.

FIG. 9 is a simplified diagram illustrating a small signal model for a Frac-N PLL according to embodiments of the present invention.

FIG. 10 is a simplified diagram illustrating a digital high-pass filter according to embodiments of the present invention.

FIG. 11 provides plots showing simulated CDR error transfer function peaking (top) and output jitter (bottom) as function of Frac-N PLL bandwidth with digital high-pass filter.

FIG. 12 is a simplified diagram illustrating a small signal model for a Frac-N PLL with charge pump according to embodiments of the present invention.

FIG. 13 is a simplified diagram illustrating a Frac-N PLL implementation with a charge pump according to embodiments of the present invention.

FIG. 14 shows the simulated CDR error transfer function peaking (top) and output jitter (bottom) as function of Frac-N PLL bandwidth with charge pump in the feedforward path.

FIG. 15 is a simplified small signal model for a Frac-N PLL with feed-forward VCO path according to embodiments of the present invention.

FIG. 16 is a simplified diagram illustrating a Frac-N PLL implemented with feed-forward VCO path according to embodiments of the present invention.

FIG. 17 shows the simulated CDR error transfer function peaking (top) and output jitter (bottom) as function of Frac-N PLL bandwidth with a feed-forward VCO path.

FIG. 18 shows the effect of gain mismatch on CDR error transfer function peaking.

FIG. 19 is a simplified block diagram illustrating feedforward gain calibration for Frac-N PLL implementation of CDR according to embodiments of the present invention.

FIG. 20 shows transfer functions for each of the two paths. It is to be noted that signal injected in Frac-N PLL path at frequency F₂ has the same gain compared to the signal injected at frequency F₁ in the feedforward path.

FIG. 21 is a simplified diagram illustrating a feedforward configuration with injected signals according to embodiments of the present invention.

FIG. 22 provides simulation results of a feedforward implementation according to embodiments of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

The present invention relates to data communication and electrical circuits. More specifically, embodiments of the present invention provide a clock and data recovery (CDR) architecture implementation for high data rate wireline communication links. In an embodiment, a CDR device includes a phase detector, a loop filter, and a fractional-N PLL. The fractional-N PLL generates output clock signal based on output of the loop filter. There are other embodiments as well.

Low-jitter CDR architectures are often an essential aspect for high data rate wireline receivers. While some of the early CDRs utilized analog voltage-controlled oscillator (VCO) based architectures, phase interpolator (PI)-based clock data recovery (CDR) modules have come to dominate CDR loop implementations due to their relative small-area (as compared to other designs and architectures) and digital control functionality. A shift towards analog-to-digital converter (ADC) based receivers (thus moving from analog to digital) has made digital control necessary for CDR loop implementations. Phase interpolators provide this functionality by converting digital phase control input signal to analog clock phase shift signal at the output. Performance of a PI in a CDR implementation requires high linearity, high output phase resolution, and high-clock-frequency operation. Unfortunately, designing highly linear PIs with high output phase resolution at high clock frequency is difficult.

As a result, CDR architectures utilizing PIs struggle to achieve the low-jitter performance criteria needed for high data rate operation. An alternative to PI-based CDR is a digitally controlled oscillator (DCO) based CDR architecture. The low-jitter requirement mentioned above precludes the use of ring oscillator based DCOs applications that require high data rates. For example, LC-based DCO (LC DCO) can potentially provide low jitter, but they are susceptible to electromagnetic coupling due to low CDR bandwidth achieved by typical high data rate links.

It is thus to be appreciated that embodiments of the present invention provide high-performance CDRs implemented with fractional-N (Frac-N) phase-lock loops (PLL). More specifically, embodiments of the present invention provide CDRs with wide bandwidth Frac-N PLLs to provide the functionalities of digitally controlled oscillators and reduce susceptibility of LC-VCOs to electromagnetic coupling, while providing low jitter digital phase shift capability to CDR loops. In an exemplary embodiment, a CDR architecture is designed to eliminate the need for phase interpolator (PI) by employing fractional-N phase-locked loop (PLL), and it is capable of achieving low-jitter performance critical for high-data rate applications. The present invention also provides methods to ensure CDR loop stability. The present invention additionally provides calibration technique to ensure robust operation. As an example, the following techniques are provided:

-   -   Use of fractional-N (analog or digital) PLLs in CDR loops for         phase and frequency tracking;     -   Digital high-pass filter based technique for compensating         adverse effects (such as stability and error transfer function         peaking) caused due to introduction of Frac-N PLL in CDR loop;     -   Charge pump feedforward techniques for compensating adverse         effects (such as stability and error transfer function peaking)         caused due to introduction of Frac-N PLL in CDR loop;         -   A variation of this technique can be used for digital             fractional-N PLLs;     -   VCO feedforward techniques for compensating adverse effects         (such as stability and error transfer function peaking) caused         due to introduction of an analog or digital Frac-N PLL in CDR         loop;     -   VCO feedforward gain calibration techniques for in-situ         calibration of parallel path gain mismatch in VCO feedforward         scheme.

The following description is presented to enable one of ordinary skill in the art to make and use the invention and to incorporate it in the context of particular applications. Various modifications, as well as a variety of uses in different applications will be readily apparent to those skilled in the art, and the general principles defined herein may be applied to a wide range of embodiments. Thus, the present invention is not intended to be limited to the embodiments presented, but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

In the following detailed description, numerous specific details are set forth in order to provide a more thorough understanding of the present invention. However, it will be apparent to one skilled in the art that the present invention may be practiced without necessarily being limited to these specific details. In other instances, well-known structures and devices are shown in block diagram form, rather than in detail, in order to avoid obscuring the present invention.

The reader's attention is directed to all papers and documents which are filed concurrently with this specification and which are open to public inspection with this specification, and the contents of all such papers and documents are incorporated herein by reference. All the features disclosed in this specification, (including any accompanying claims, abstract, and drawings) may be replaced by alternative features serving the same, equivalent or similar purpose, unless expressly stated otherwise. Thus, unless expressly stated otherwise, each feature disclosed is one example only of a generic series of equivalent or similar features.

Furthermore, any element in a claim that does not explicitly state “means for” performing a specified function, or “step for” performing a specific function, is not to be interpreted as a “means” or “step” clause as specified in 35 U.S.C. Section 112, Paragraph 6. In particular, the use of “step of” or “act of” in the Claims herein is not intended to invoke the provisions of 35 U.S.C. 112, Paragraph 6.

Please note, if used, the labels left, right, front, back, top, bottom, forward, reverse, clockwise and counter clockwise have been used for convenience purposes only and are not intended to imply any particular fixed direction. Instead, they are used to reflect relative locations and/or directions between various portions of an object.

FIG. 1 is a simplified diagram illustrating a fractional-N PLL according to an embodiment of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. The PLL includes a phase detector (PD) or phase and frequency detector (PFD), a charge pump (CP), an analog loop filter (ALF), a voltage-controlled oscillator (VCO), multi-modulus divider (MMD), and a digital ΔΣ modulator (ΔΣ). The input reference clock is denoted as CLK_(REF), and the output clock is denoted as CLK_(OUT). A high resolution digital frequency control signal, D_(CTRL), is converted to an integer division ratio, N_(DIV), by a digital ΔΣ modulator. The output of the multi-modulus divider (MMD) is divided clock signal denoted as CLK_(DIV). The output frequency, F_(OUT), of such a fractional-N PLL depends on the input reference frequency F_(REF) and D_(CTRL), which is described in Equation 1 below: F_(OUT)=D_(CTRL)F_(REF)  Equation 1:

It is to be noted that D_(CTRL) can be used to control not only the output frequency, but also the output phase of the Frac-N PLL. FIG. 2 provides a small signal model of a multi-modulus divider according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. It can be shown that the overall output phase of the MMD is given by Equation 2 below:

$\begin{matrix} {{\Phi_{DIV}(z)} = {{N_{nom}{\Phi_{OUT}(z)}} + {2{\pi\left( {{N_{DIV}(z)} - N_{nom}} \right)}\left( \frac{z^{- 1}}{1 - z^{- 1}} \right)}}} & {{Equation}\mspace{14mu} 2} \end{matrix}$

where Φ_(DIV) is the phase of the divider output, Φ_(OUT) is the phase of the fractional-N PLL output that is input to the MMD, N_(nom) denotes the nominal division ratio of the fractional-N PLL, and N_(DIV) denotes the instantaneous integer division ratio of MMD.

Equation 2 can be expressed as Equation 3:

$\begin{matrix} {{\Phi_{DIV}(z)} = {{N_{nom}{\Phi_{OUT}(z)}} + {2{\pi\left( {{{D_{CTRL}(z)}ST{F(z)}} - N_{nom}} \right)}\left( \frac{z^{- 1}}{1 - z^{- 1}} \right)}}} & {{Equation}\mspace{14mu} 3} \end{matrix}$

where STF(z) is the signal transfer function of the digital ΔΣ modulator and D_(CTRL) denotes the digital frequency control input. If we assume an ideal reference clock for the Frac-N PLL (i.e., Φ_(DIV)=0), the output phase is given by Equation 4 below:

$\begin{matrix} {{\Phi_{OUT}(z)} = {{- \frac{2\pi}{N_{nom}}}\left( {{{D_{CTRL}(z)}ST{F(z)}} - N_{nom}} \right)\left( \frac{z^{- 1}}{1 - z^{- 1}} \right)}} & {{Equation}\mspace{14mu} 4} \end{matrix}$

Equation 4 shows that the output phase of the Frac-N PLL can be controlled using digital frequency control signal, D_(CTRL).

FIG. 3 is a simplified diagram illustrating a DCO-based CDR loop. The phase difference between incoming data, DIN, and sampling clock CLK_(OUT) is detected by a digital phase detector (Digital PD). As an example, a proportional-integral path based digital loop filter (Digital LF) is used to generate a digital frequency control word D_(CTRL) for the DCO. The output frequency F_(OUT) of the DCO depends on D_(CTRL), and the output frequency (F_(OUT)) of the DCO (characterized by a DCO gain, K_(DCO)) is described in Equation 5 below: F_(OUT)=D_(CTRL)K_(DCO)  Equation 5:

It is to be appreciated that there are similarities between DCO and Frac-N PLL. In particular, the functionality of Frac-N PLL in CDR implementations according to embodiments of the present invention is equivalent to a digitally controlled oscillator (DCO), with the functional characteristics K_(DCO)=F_(REF). In various embodiments, a Frac-N PLL is used as a functional equivalent of a DCO in a CDR loop. Compared to a stand-alone LC DCO implementation, a wide bandwidth Frac-N PLL provides better immunity to VCO coupling noise. Due to its closed loop operation, it also provides a DCO implementation with a well-defined process independent DCO gain, K_(DCO).

FIG. 4 is a simplified diagram illustrating a Frac-N PLL incorporated inside a CDR loop according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. The digital frequency control word D_(CTRL) (generated for CDR loop) is used as a frequency control input for a Frac-N PLL. The output clock generated by Frac-N PLL, CLK_(OUT), is used as a sampling clock for the CDR loop.

As explained above, Frac-N PLL in FIG. 4 provides functionalities and advantages of an LC DCO without its penalties as used in a CDR. As an example, Frac-N PLL includes a phase detector (PD) or a phase-frequency detector (PFD) for phase detection, a charge pump for compensation, an analog loop filter (ALF) for filtering. The Frac-N PLL additionally includes an oscillator, an MMD, and a digital ΔΣ modulator. It is to be appreciated that Frac-N PLL can be implemented in other ways as well. It is to be understood CDRs according to the present invention can be implemented using analog Frac-N PLLs or digital Frac-N PLLs.

The incorporation of a Frac-N PLL in a CDR also contributes to certain functional difference from an LC DCO implementation. After all, a Frac-N PLL is not the same as an LC DCO. More specifically, while a Frac-N PLL has well-defined gain at low frequencies, its gain rolls off at high frequencies due to low pass nature of the PLL. FIG. 5 is a simplified diagram illustrating small signal phase domain model of a CDR loop with a DCO.

In case of an ideal DCO, the gain is independent of frequency, (i.e., K_(DCO)(s)=K_(DCO), where s is the reference modulation frequency). The digital loop filter is chosen to achieve appropriate CDR closed loop bandwidth and jitter tolerance. On the other hand, when Frac-N PLL is used (instead of DCO), the gain is given by K_(DCO)(s)=F_(REF)G(s), where G(s) is effective low-pass transfer function of the Frac-N PLL. Due to its low-pass nature, G(s) introduces additional poles in the CDR loop transfer function, which adversely affects the CDR loop stability and creates peaking in the CDR error transfer function E(s), where E(s) is defined by E(s)=Φ_(E)(s)/Φ_(IN)(s), subsequently degrading the jitter tolerance of the CDR loop. Note that jitter tolerance and error transfer function are related as JTOL(s)=1/E(s).

FIG. 6 is a simplified diagram illustrating a small signal model of Frac-N PLL functioning as a DCO according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. The function F_(OUT)(s) represents the output frequency of the Frac-N PLL, while Φ_(OUT)(s) represents its phase. The high-pass shaped quantization noise of the digital ΔΣ modulator is denoted as E_(Q,NTF)(s). The behavior of the Frac-N PLL is described in Equation 6 below:

$\begin{matrix} {{\frac{F_{OUT}(s)}{D_{CTRL}(s)} = {\frac{F_{OUT}(s)}{E_{Q,{NTF}}(s)} = {F_{REF}{G(s)}\mspace{14mu}{Where}}}},{{G(s)} = \frac{L{G(s)}}{1 + {L{G(s)}}}}} & {{Equation}\mspace{14mu} 6} \end{matrix}$

In Equation 6, LG(s) denotes the open loop gain of the Frac-N PLL. To reduce CDR error transfer function E(s) peaking, the implementation needs to increase the bandwidth of Frac-N PLL and by extending the bandwidth of G(s). This increased bandwidth, however, allows more noise from digital ΔΣ pass through. As a result, there is a trade-off between peaking of E(s) and the output jitter of the Frac-N PLL. FIG. 7 is a plot illustrating CDR error transfer functions comparing ideal DCO and actual DCO implementation (e.g., with Frac-N PLL) with a pole that contributes to undesired error transfer function peaking.

FIG. 8 provides plots illustrating CDR error transfer function peaking (top) and Frac-N PLL jitter (bottom) as function of Frac-N PLL bandwidth. As can be seen in FIG. 8, the peaking is reduced at higher PLL bandwidth at the cost of worse jitter performance. For optimal system performance, the implementation needs to reduce the bandwidth for F_(OUT)(s)/E_(Q,NTF)(s) while maximizing the bandwidth for F_(OUT)(s)/D_(CTRL)(s).

In various embodiments, the present invention provides feedforward techniques to compensate the bandwidth limitations of Frac-N PLL. The problem discussed in the previous section can be solved by introducing a feedforward path from D_(CTRL)(s) to F_(OUT)(s). Such a feedforward path can decouple the transfer function seen by D_(CTRL)(s) from the transfer function seen by E_(Q,NTF)(s). We propose the following methods for implementing the feedforward path.

According to a specific embodiment, a Frac-N PLL is implemented in a CDR with a digital high-pass filter. FIG. 9 is a simplified diagram illustrating a small signal model for a Frac-N PLL according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. It is to be appreciated that the addition of a digital high-pass filter in the D_(CTRL) path can effectively counter the low-pass nature of G(s). In this case, the DCO (i.e., DCO equivalent implemented with Frac-N PLL as used in a CDR) transfer function is given by Equation 7 below:

$\begin{matrix} {\frac{F_{OUT}(s)}{D_{CTRL}(s)} = {F_{REF}{H_{1}(s)}{G(s)}}} & {{Equation}\mspace{14mu} 7} \end{matrix}$

It is to be noted that the transfer function from E_(Q,NTF) to F_(OUT) remains unchanged. For example, for H₁(s)=G⁻¹(s), Equation 8 below is obtained:

$\begin{matrix} {\frac{F_{OUT}(s)}{D_{CTRL}(s)} = {{F_{REF}{G^{- 1}(s)}{G(s)}} = F_{REF}}} & {{Equation}\mspace{14mu} 8} \end{matrix}$

In practical implementations, a digital finite impulse response (FIR) filter can be used to approximately match the inverse of G(s). Furthermore, there is a possibility of noise in D_(CTRL) signal folding back in Frac-N PLL in-band frequency region due to up-sampling in the Frac-N PLL. To alleviate this undesirable effect of out of band noise, an anti-aliasing filter is also provided in the D_(CTRL) path. This anti-aliasing filter can also be implemented as a digital FIR filter. FIG. 10 is a simplified diagram illustrating a digital high-pass filter according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. For example, a digital high pass filter H₁(s) includes an inverse filter H₁(z) and an anti-aliasing filter AA(z).

FIG. 11 provides plots showing simulated CDR error transfer function peaking (top) and output jitter (bottom) as function of Frac-N PLL bandwidth with digital high-pass filter. It can be seen that the CDR error transfer function peaking becomes almost independent of Frac-N PLL bandwidth. This allows for a Frac-N PLL bandwidth that optimizes the jitter performance.

In certain embodiments, a CDR is implemented with Frac-N PLL and a charge pump, where the charge part is configured on the feedforward path. FIG. 12 is a simplified diagram illustrating a small signal model for a Frac-N PLL with charge pump according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. The DCO (i.e., DCO equivalent implemented with Frac-N PLL as used in a CDR) transfer function is provided in Equation 9 below:

$\begin{matrix} {\frac{F_{OUT}(s)}{D_{CTRL}(s)} = {{F_{REF}{G(s)}} + {\frac{sN_{nom}}{\alpha\; I_{CP}}{H_{2}(s)}{G(s)}}}} & {{Equation}\mspace{14mu} 9} \end{matrix}$

where α and I_(CP) denote phase detector and charge pump gain, respectively. It is to be noted that the transfer function from E_(Q,NTF) to F_(OUT) remains unchanged. If we choose

${{H_{2}(s)} = {\frac{\alpha\; I_{CP}F_{REF}}{sN_{nom}}\left( {{G^{- 1}(s)} - 1} \right)}},$ Equation 10 below is obtained:

$\begin{matrix} {\frac{F_{OUT}(s)}{D_{CTRL}(s)} = F_{REF}} & {{Equation}\mspace{14mu} 10} \end{matrix}$

This feedforward path implementation requires digital filter as well as a current mode ΔΣ digital-to-analog converter (IDAC). FIG. 13 is a simplified diagram illustrating a Frac-N PLL implementation with a charge pump according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. The IDAC is configured with ΔΣ modulator in the signal path as shown. It is to be understood while PD and CP are shown in the same block in FIG. 13, they are implemented separately.

FIG. 14 shows the simulated CDR error transfer function peaking (top) and output jitter (bottom) as function of Frac-N PLL bandwidth with charge pump in the feedforward path. It can be seen that the CDR error transfer function peaking becomes almost independent of Frac-N PLL bandwidth. This allows for choosing Frac-N PLL bandwidth that optimizes the jitter performance.

In various embodiments, a Frac-N PLL is implemented with a feed-forward VCO path as a part of a CDR device. FIG. 15 is a simplified small signal model for a Frac-N PLL with a feed-forward VCO path according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. As shown in FIG. 15, the VCO (with gain K_(VCO)) is configured on the feedforward path. The DCO (i.e., DCO equivalent implemented with Frac-N PLL as used in a CDR) transfer function is described in Equation 11 below:

$\begin{matrix} {\frac{F_{OUT}(s)}{D_{CTRL}(s)} = {{F_{REF}{G(s)}} + {\frac{sN_{nom}}{\alpha\; I_{CP}{H_{LPF}(s)}}{H_{3}(s)}{G(s)}}}} & {{Equation}\mspace{14mu} 11} \end{matrix}$

where H_(LPF)(s) denotes the equivalent low pass transfer function of the analog loop filter connected to charge pump output. It is to be noted that the transfer function from E_(Q,NTF) to F_(OUT) remains unchanged. If we choose

${{H_{3}(s)} = \frac{F_{REF}}{K_{VCO}}},$ Equation 12 below is obtained:

$\begin{matrix} {\frac{F_{OUT}(s)}{D_{CTRL}(s)} = F_{REF}} & {{Equation}\mspace{14mu} 12} \end{matrix}$

Since H₃(s) is a constant, this feedforward path implementation requires only a ΔΣ digital-to-analog converter. FIG. 16 is a simplified diagram illustrating a Frac-N PLL implemented with a feed-forward VCO path according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. For example, the ΔΣ digital-to-analog converter can be implemented as a switching capacitor in the VCO as shown in FIG. 16. For example, a digital gain scaling factor K_(VFF) is utilized to implement H₃(s) function.

FIG. 17 shows the simulated CDR error transfer function peaking (top) and output jitter (bottom) as function of Frac-N PLL bandwidth with a VCO on the feedforward path. It can be seen that the CDR error transfer function peaking becomes almost independent of Frac-N PLL bandwidth. This allows for choosing a Frac-N PLL bandwidth that optimizes the jitter performance.

It is to be noted that a VCO contributes to Frac-N PLL feedforward gain variation. As discussed before, the feedforward gain must match with Frac-N PLL K_(VCO) to achieve an all-pass transfer function from D_(CTRL) to F_(OUT). This gain scaling is implemented in digital domain, whereas the K_(VCO) of the analog VCO varies with process, temperature, and supply voltage. As a result, there is a mismatch between the digital gain scaling factor (K_(VFF)) and the Frac-N PLL gain. FIG. 18 shows the effect of gain mismatch on CDR error transfer function peaking. It indicates that for optimum performance, the digital path gain K_(VFF) needs to be calibrated to match the analog gain inside Frac-N PLL.

FIG. 19 is a simplified block diagram illustrating feedforward gain calibration for Frac-N PLL implementation of CDR according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. More specifically, the foreground calibration technique illustrated in FIG. 9 involves matching digital feedforward path gain with that of analog path. As discussed above, the gain in the feedforward path (e.g., the H_(VFF)(s) function) is to be matched with the gain Frac-N PLL path (e.g., the H_(FRAC)(s) function).

FIG. 20 shows transfer functions for each of the two paths. It is to be noted that signal injected in Frac-N PLL path at frequency F₂ has the same gain compared to the signal injected at frequency F₁ in the feedforward path. Variables F_(CDR) and F_(PLL) denote the bandwidths for CDR and Frac-N PLL respectively. To measure the output because of the injected signal, we utilize the CDR phase detector. Since the CDR phase detector provides digital output, it can be measured in-situ.

FIG. 21 is a simplified diagram illustrating a feedforward configuration with injected signals according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. Digital square wave signals are injected through each path, and they create triangular signals at the CDR phase detector output. By detecting the peak-to-peak value of the triangular waveforms digitally, we can find the gain mismatch and correct for it by digitally adjusting K_(VFF). FIG. 22 provides simulation results of a feedforward implementation according to embodiments of the present invention. As an example, it shows that for an expected ratio 10 for peak to peak values, K_(VFF) path gain needs to match with Frac-N PLL path gain. For other scaling factors the ratio of peak-to-peak values deviates from 10.

It is to be appreciated that Frac-N PLLs can be implemented into CDR devices in various ways. For example, the combination and configuration of charge pump, VCO, and/or other components of a Frac-N PLL can be used to allow the Frac-N PLL to satisfy the requirements of CDR devices.

While the above is a full description of the specific embodiments, various modifications, alternative constructions and equivalents may be used. Therefore, the above description and illustrations should not be taken as limiting the scope of the present invention which is defined by the appended claims. 

What is claimed is:
 1. A clock data recovery (CDR) device comprising: a digital phase detector circuit configured to detect a phase difference of an incoming data signal and to generate a first phase detection signal using a sampling clock disposed on a feedback path; a digital loop filter configured to generate a frequency control word using at least the first phase detection signal; a feedforward path circuit comprising a first sigma delta modulator, the first sigma delta modulator being configured to generate an oscillator compensation signal according to at least the frequency control word and a digital gain scaling factor; and a fractional-N phase lock loop (Frac-N PLL) circuit comprising a voltage-controlled oscillator (VCO) configured to generate the sampling clock signal using an oscillator control signal and the oscillator compensation signal, the oscillator control signal being based on a reference clock signal and the frequency control word, the oscillator compensation signal being generated to reduce a frequency roll off of the sampling clock signal introduced by the Frac-N PLL circuit.
 2. The CDR device of claim 1, wherein the Frac-N PLL circuit comprises: a second sigma delta modulator configured to generate a frequency control signal according to the frequency control word; a frequency divider circuit configured to generate a divided clock signal according to at least the sampling clock signal and the frequency control signal; and a phase detector configured to generate a second phase detection signal according to the reference clock signal and the divided clock signal.
 3. The CDR device of claim 2, wherein the Frac-N PLL circuit further comprises: a charge pump configured to generate intermediate current pulses based on the second phase detection signal; and a loop filter configured to generate the oscillator control signal according to the intermediate current pulses.
 4. The CDR device of claim 2, wherein the second sigma delta modulator is configured to convert the frequency control word to the frequency control signal, the frequency control signal being based on an integer division ratio.
 5. The CDR device of claim 2, wherein the frequency divider circuit comprises a multi-modulus divider configured to generate the divided clock signal according to at least the sampling clock signal and the frequency control signal.
 6. A method for performing clock data recovery, the method comprising: receiving an incoming data signals; detecting a phase difference using the incoming data signal and a sampling clock signal, the sampling clock signal being configured on a feedback path; generating a first phase detection signal based on the phase difference; generating a frequency control word based on the first phase detection signal using a digital loop filter; generating an oscillator compensation signal to compensate a high-frequency roll off according to at least the frequency control word and a digital gain scaling factor by a feedforward path circuit, the feedforward path circuit comprising a first sigma delta circuit; receiving a reference clock signal; generating an oscillator control signal based at least on the reference clock signal and the frequency control word by a fractional-N phase lock loop (Frac-N PLL) circuit, the Frac-N PLL comprising a voltage controlled oscillator (VCO); and generating the sampling clock signal using at least the oscillator control signal and the oscillator compensation signal by the VCO.
 7. The method of claim 6 further comprising: generating an integer division ratio according to the frequency control word; and generating an intermediate clock signal based on at least the sampling clock signal and the integer division ratio.
 8. The method of claim 7 further comprising: generating a second phase detection signal according to the reference clock signal and the intermediate clock signal; and generating the oscillator control signal, by an analog loop filter, according to at least the second phase detection signal.
 9. The method of claim 8 further comprising generating a current pulse according to the second phase detection signal, the current pulse being used by the analog loop filter to generate the oscillator control signal.
 10. The method of claim 6 further comprising calculating the digital gain scaling factor according to at least the reference clock signal and the sampling clock signal, the digital gain scaling factor being a function of a ratio between the reference clock signal and the sampling clock signal. 